Sync Pulse Separation and Use [PHOTOFACT Television Course (1949)]

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Up to this point, we have considered those sections of a television receiver concerned with the formation of an electron beam or "pencil" in a cathode-ray picture tube, the control of this beam in intensity, and its deflection in the horizontal and vertical directions.

It has been shown that the path traced by the beam can be made to cover the picture area in a regular sequence, which corresponds to a similar sequential scanning action occurring in the "pick-up or camera" tube at the transmitter. The methods by which synchronized control of the receiver scanning action is accomplished have been described.

In the following sections we will cover those portions of the receiver which concern:

1. The interception and selection of the desired television signal and its amplification.

2. The detection and separation of the video or "sight" portion of the signal from the accompanying sound signal.

3. The use of the video signal to modulate the intensity of the picture tube electron beam.

4. The reconstruction or insertion of the average or "background light level" (de-restoration).

5. The separation, amplification and use of the sync pulses.


Sync pulse separation has been mentioned in connection with the description of deflection systems. At this time we will consider in greater detail the methods of separation, amplification and subsequent use of the sync pulses. Before starting such a study, it will be of value to review as a "re fresher", some of the fundamental actions and characteristics of vacuum tubes as amplifiers and rectifiers.

REVIEW OF THE ACTION OF VACUUM TUBES AS APPLIED TO THE SEPARATION AND USE OF SYNC PULSES: In analyzing commercial receivers with regard to the method of separating the sync pulses from the "composite" video signal, we find circuits in which this function is accomplished by diodes, triodes and pentodes employed under widely different circuit and operating conditions.

These include: diode and inverted diode rectification, triodes and pent odes working at either plate current cut-off or saturation, and "cathode follower" circuits. As an introduction to our study of the individual applications and their use in typical television circuits, the following review of vacuum tube action should be considered together with a study of the characteristic curve s and data presented in receiving tube manuals available from the tube manufacturers.

1. The limits of the plate current range of a vacuum tube are determined by cut-off and saturation. Plate current cut-off can be accomplished by a fixed negative grid bias or by negative bias derived from grid circuit rectification of the signal in the case of triodes and pent odes. A series "delay" bias can produce cut-off in the case of diodes. Plate current saturation is produced by operation at low plate voltage in the case of triodes, and at low plate and screen voltages with pentodes.

2. The voltage developed across a cathode resistor, by the plate current flow, pro vides a negative bias if the grid circuit is returned to the end of the resistor opposite the cathode. A tube biased by its own plate current cannot cut itself off. Self bias from the cathode resistor is often augmented by additional negative bias in series with the grid return or by "bleeding" current through the cathode resistor from the plate supply, to assure operation at or beyond cut-off. (See Figure 91.)

3. A change of the control grid voltage will produce a change of plate current in the same direction or polarity. Stated in another way, the grid voltage and the plate current are in phase with one another.

Fig. 91. Cut-off by Combination Bias

4. The plate voltage change caused by a change of grid voltage is in the opposite direction and hence the plate voltage is 180° out of phase with the grid voltage. For this reason, a vacuum tube amplifier is essentially a phase inverter.

5. A drop of plate voltage caused by a positive pulse applied to the grid is known as a "negative going" pulse. As we have seen, such a pulse is required for the control of a cathode-coupled multivibrator.

6. An increase in plate voltage caused by a pulse, which drives the grid more negative than its initial bias condition, is known as a "positive going" pulse. This is the requirement for "tripping" a blocking oscillator.

7. The effect of a plate load resistor is to cause the plate-cathode voltage to decrease with an increase in plate current. Conversely, with such a load, the plate to cathode voltage will increase with a drop in plate current.

8. The amount of change in plate to cathode voltage caused by a change in plate current can be transmitted or transferred to the grid circuit of a following tube, in the same polarity, by means of a coupling capacitor.

9. When the load resistor of a vacuum tube is in the cathode to "B" minus circuit of the tube and the output is taken from the cathode (see Figure 93) the circuit is known as a "cathode follower". Since this type of circuit is frequently employed in television receivers and has not been widely used in broadcast receivers, its action will be described before considering the general subject of sync separation.

Fig. 92. The Basic Cathode Follower Circuit

Fig. 93. Demodulation (Detection) of the Video (Picture) Carrier

THE CATHODE FOLLOWER CIRCUIT: Figure 92 shows a circuit having many interesting characteristics. Its features are: a high input impedance, a low input capacitance, a low output impedance, and an output voltage which

''follows" the input voltage and hence is of the same polarity. Although it is a power amplifier, the voltage gain can never reach unity and therefore the loss which it introduces often has to be compensated by additional amplification.

These characteristics result from the fact that the unbypassed cathode load resistor (R2) produces inverse or negative feedback to the grid.

The action of the circuit is as follows:

1. The plate bypass capacitor (C2) is made sufficiently large that, for the lowest frequency to be encountered, the plate is substantially at ground potential. The cathode resistor (R2) is unbypassed or if a bypass capacitor is used it is made of low value so that it s reactance is high at the highest frequency which the stage is required to pass.

The grid is biased negative with respect to cathode by the plate current which flows through the cathode resistor (R2). This automatically establishes an operating point near, but not at, plate current cut-off.

2. Under operating conditions just de scribed, let us examine the results of applying a positive pulse or signal to the grid. The plate current rises, and the increased plate current flowing through R2 causes the cathode to become more positive. If a negative pulse is applied to the grid, the consequent drop in plate current will decrease the voltage drop across R2, and cause the cathode to become less positive with respect to ground. Since the voltage change of the cathode in each case is in the same direction as the grid change which causes it, the cathode "follows" the grid and the circuit is consequently known as a cathode follower.

3. The effective control voltage of the tube, or voltage between grid and cathode, is the difference between the input voltage across R1 and the voltage across the cathode resistor (R2). Thus, the variation of cathode voltage caused by the varying input decreases the effectiveness of the input signal and the stage is said to be "degenerative".

4. This "degeneration" or negative feedback produces the effects of low loading on any circuit across which the stage is connected (high input resistance and low input capacitance). It is interesting to note that the effective input capacitance can be lower than the measured value with the tube cold. To under stand how this seemingly impossible condition can occur, let us examine the manner in which the tube and circuit can change the voltage conditions as viewed from the input terminals.

5. Let us assume that when, with a particular tube and plate voltage, a voltage change of 1 volt appears across the input terminals and that this impressed voltage causes a resultant change of cathode to ground voltage of .9 volts. The applied input voltage "attempts" to charge the input capacitance of the tube to a potential of 1 volt. Immediately, there appears a voltage of the same polarity from the cathode on the other "plate" of this capacitor and the resultant charge is 0.1 volt rather than 1 volt.

In this way, the input capacitance of the tube is apparently much lower than its actual "static" value. A similar effect occurs with respect to the input resistance of the tube. In the example used, the effective input capacitance would be only one-tenth of the actual capacitance.

6. In a similar fashion, the effective out put impedance of the stage is very low. Its value depends upon the amplification factor and the mutual or transconductance of the tube. As an example, a tube such as the 6J5 with a mu of 20, a plate resistance of 7700 ohms and a Gm (transconductance) of 2600 micro-mhos, will have an effective output impedance across a 1000 ohm cathode load resistor of only 268 ohms.

The cathode follower, which was used extensively in radar equipment during the war, finds many applications as a coupling means in television transmitters and receivers.

SYNCHRONIZING PULSE SEPARATION: The nature of the horizontal and vertical synchronizing pulses and their time relationship to the sawtooth scanning raster were described and illustrated here. It was indicated that these pulses occurred during the time that the electron stream in the picture tube is cut off or "blanked" out. While we will examine all of the details of the complex television signal later in the course, it will be instructive at this time to examine the video IF signal as it reaches the video detector. The manner in which the sync pulses are separated from the signal and used to control the horizontal and vertical scanning systems of the receiver will be discussed. Since the horizontal pulses occurring at the end of each of the 525 lines which constitute the picture, are representative of the pulse separation principle, our discussion at present will be confined to their separation and utilization. A treatment of the vertical pulses will be covered when we consider the methods of segregating the vertical from the horizontal pulses.

The pulses can be clipped from the signal in three places in the circuit:

1. At the input of the video or "sight" detector.

2. From any of the video amplifying stages.

3. At the point of restoration of the average background light of the picture. For this reason, Figure 93 shows the nature of the video signal with its picture and synchronizing "in formation" as it appears at the output of the last video IF amplifier. The video IF carrier at the input of the video second detector, shown in Figure 93 as a diode, is represented at point ''A''. It will be seen to consist of horizontal pulses (as detailed in Figure 65) mounted on a ''shelf'' or ''pedestal''. These are identified in Figure 93 as (1), the pulse, and (2), the pedestal. Between the edges of the pedestals is found the modulated envelope of the carrier which represents the variations of light and dark of the video signal (3), used to modulate the cathode ray beam.

Figure 93B shows the form of the detected or demodulated wave with the sync pulses in the region from ( 4) to ( 5). The "picture information" or video signal is shown in the portion of the wave shown at ( 6). It will be noted that the sync pulses are at the "top" of the signal. Since these occur when the picture tube is ''black'' or blanked, it will be evident that the darker tones of the picture are just below the ''pedestal'', or point (4), of Figure 93B. While the exact nature of the video signal will be discussed in detail later, it is of importance to note that, after passing through the video amplifier, the signal must reach the grid of the "picture" tube in such a phase that the sync pulses are the most negative part of the wave. This accomplishes blanking during the return trace.

We have seen that the polarity of the sync pulse, as it arrives at the grid of the scanning oscillator, must be of the proper sign to assure control. For this reason, the method of sync pulse separation selected in any particular receiver design will depend upon: the number of stages of video amplification, the point in the circuit at which sync separation is accomplished, and the number of sync amplifier or clipping stages employed.

All of the methods of separating the synchronizing pulse from the rest of the video signal, involve the fact that the "pedestal" or blanking level is always maintained in trans mission, at a definite point on the carrier wave (75% of maximum carrier) and therefore the sync pulses occupy the top 25% of the wave.

The problem of sync clipping thus resolves itself into one of '' amplitude separation'' or of removing the top 25% of the wave without passing the lower values which contain the video signal. The methods commonly employed will be covered under headings of the types of tubes used.

DIODE SYNC SEPARATION CIRCUITS. Figure 94 shows the commonly used diode sync separation circuits. The first three of these (A, B and C) involve the "clipping" of the pulse by rectification of the video IF carrier.

In each case, the diode will be found connected across the video detector circuit. The video detectors have been omitted for simplicity.

In Figure 94A the picture signal is of positive polarity, meaning that the output of the detector becomes more positive as the picture carrier increases. Since the sync pulses are at the "top" of the carrier modulation, the output from the diode separator is in the form of positive pulses. One stage, or any odd number of stages, of video amplification, which will invert the phase, will be required following a detector and separator of this type. This polarity of the video signal is correct for control of the picture tube grid, since a negative voltage is required for beam cut-off. The negative bias shown in series with the circuit of Figure 94A biases the diode to the "pedestal". Anode current flow s for the duration of the sync pulses only.

Figure 94B shows an "inverted" version of the same circuit. In this case, an increase of video carrier causes a more negative out put voltage. The pulse output is negative and the video output of the detector diode, connected across the separator, is in the correct direction to actuate the picture tube grid. For this reason, an even number of video stages must be employed, causing the output to be of the same phase as the input. As in the preceding case (Figure 94A), a delay bias is employ ed. Diode rectification is limited to the level of the sync pulses and the rest of the video signal is rejected. The bias in this case, however, is positive with respect to ground or chassis.

Fig. 94. Diode Sync Separation Circuits.

The operation of the circuit (Figure 94C) is similar to that of Figure 94A except that the bias required for delay of the diode action to the correct point is derived from rectification of the signal itself. Two load circuits are employed in series as the cathode return circuit.

A bias voltage appears across one of these ( R1-C1), and the output sync pulse is developed across the other (R2). The bias circuit ( R1-C1) features the use of a time constant which is long compared with the horizontal and vertical scanning times. The resistor is in the megohm range and the capacitor in the order of 0.5 mfd.

The D-C component resulting from the charge of the rectified sync pulses, in capacitor C1, automatically biases the diode to the "pedestal" or "black" level This type of circuit can also be used with a rectified video signal as well as with video carrier input as shown.

In the circuit of Figure 940, the bias for separation of the sync pulse from the rest of the video signal is also derived by rectification of the signal itself. The composite video signal is applied to the input terminals in such a direction that an increase of the signal tends to drive the plate of the diode more positive.

This action charges the capacitor C1 and its discharge through resistor R1 during the period between pulses establishes a negative voltage drop with respect to the plate. With proper choice of time constant, the diode can be automatically biased by the video signal to proper point for "clipping" of the sync pulses from the signal.

TRIODE SYNC SEPARATION CIRCUITS. In many receivers, triodes or pentodes are employed rather than the diode circuits just discussed. The choice of tube type depends upon a number of design factors involving type of sync control, polarity requirement, and point in the circuit at which separation is accomplished. In general, the use of these multi-element tubes allows some voltage gain to be realized in the separation process and, in certain of the circuits, the additional feature of "leveling" or "limiting". Three basic types of triode sync separating circuits are shown in Figure 95.

The circuit of Figure 95A makes use of grid rectification of the video signal, to bias the control grid, so that cut-off of the plate current occurs at the desired pedestal level.

This action is similar to that just discussed for the diode circuit of Figure 940. Two additional actions are found in this circuit:

1. The sync pulses are sufficiently high in magnitude to drive the grid into the positive region and the lowered grid resistance limits the input signal by loading.

2. Some amplification of the sync pulse occurs due to the amplifying properties of the triode.

In the arrangement shown in Figure 95B, the operating conditions a requite different from those of the circuit of Figure 95A. The tube is biased from an external source of volt age, through resistors R1 and R2, to such a point that it is just starting to draw grid current at the "black" or pedestal level. The input signal is inverted in polarity from that of the preceding example. In this case, the sync pulse portion of the video input signal is the most negative region. The plate voltage is made very low so that saturation of the plate current occurs at a grid voltage close to its zero value. The portion of the video input signal, more positive than the desired "clipping" level, lies in the saturation region and produces no further rise in plate current. For this reason, "limiting" or "leveling" occurs at this saturation point. The negative excursions of grid voltage, caused by the sync pulse portion of the input signal, cause a drop in plate current as shown in the wave form drawings of Figure 95B. Although the amplitude of the sync input pulses has been limited in the drawing (in order to better illustrate the circuit action), it is evident that if their amplitude is extended to beyond the grid cut-off point, limiting will also occur due to plate current cut-off and the output pulses will be uniform in size.




Fig. 95. Triode Sync Separation Circuits.

The series resistor in the grid circuit ( R1) limits the amount of grid current which can flow over any one frame. This prevents a long-time blocking condition from developing due to an excessive charge on capacitor C1.

The circuit of Figure 95C employs cathode bias to establish the correct operating point for plate circuit separation of the sync pulses. In this case, the values chosen for resistor R2 and capacitor C2 are determined by the following considerations:

1. The resistor must have such a value that the plate current pulses above the "clip ping'' level will produce a voltage-drop equal to the required operating point. For high-mu tubes, this resistor will be in the order of 10,000 ohms.

2. The value of the capacitance C2 must be a compromise between several factors. It must be high enough to maintain constant bias voltage throughout at least one vertical blanking period and yet low enough in value that it can change its charge in accordance with changes of the average background lighting of the scene.

Fig. 96. Pentode Sync Separating Circuit Incorporating "Leveling" or "Limiting"

PENTODE SYNC SEPARATION. Figure 96 illustrates the use of a sharp cut-off pentode as a combination sync separator and limiting amplifier. Plate current saturation is assured by operating both the screen and plate at extremely low voltages. This is accomplished by the use of dividing networks. The screen network consists of resistors R3 and R4 and it will be seen that with a "B" supply voltage of 340 volts, the screen is held at the extremely low voltage of 3.2 volts. In like manner, the plate supply network consisting of resistors R5 and R6 maintains the plate at 2.6 volts. Under these conditions, plate current saturation occurs just after the grid voltage has passed into the positive region as shown in the diagram of Figure 96. The grid circuit network, consisting of resistors R2 and R1 with capacitor C1 establishes the operating point of the circuit by grid circuit rectification of the video signal. This assures plate current cut-off at a level just above the "pedestal" or "black" region. It will be seen from the drawings that the synchronizing pulses are clipped at both ends and leveling or limiting occurs. Since a pent ode operating under these conditions exhibits very low volt age gain, it is necessary to operate a circuit of this type either at a high voltage level point in the system or to follow it with sync amplifiers.

The series grid resistor (R2) in this circuit functions in a manner similar to that of the positive grid triode of Figure 95B. In this case, however, it serves an additional function.

Sharp noise pulses, such as those due to motor car ignition, whose amplitude might be higher than the sync pulses in the absence of this resistor would drive the grid highly positive and result in a blocking condition. This would cause loss of synchronization, during the period that the tube was blocked, resulting in loss of grid and plate current which can flow during any one sync or noise pulse.

CATHODE FOLLOWER SYNC SEPARATION CIRCUITS. Figure 97 shows two circuits which employ the cathode follower principle for the separation of synchronizing pulses from the video signal. In both cases the action is at high voltage level and the input to the cathode follower is taken from the final stage of the video amplifier.

Figure 97A shows a 6V6 output pentode operating as a high transconductance triode with screen and plate connected together.

This tube, in turn, feeds a diode acting as a sync clipper in a manner similar to that described in connection with Figure 94B. Several unusual circuit feature s are evident:

1. The grid circuit of the cathode follower is returned to plus 300 volts through the network consisting of R1, R2 and R6. If the diode (T2) were to be removed from its socket so that no plate current could flow through R6, the grid of tube T1 would be at a positive potential.

2. The circuit consisting of resistor R6 and capacitor C4 is charged by the rectified sync pulses from tube T2. The values chosen for the components of this circuit produce a long time constant. A steady bias voltage, negative to ground, appears across R6.


Fig. 97. Cathode Follower Sync Separation Circuits.

3. This bias voltage is introduced into the grid circuit at the tap between resistors R1 and R2 and opposes the positive voltage from the plate circuit.

4. Plate current of the cathode follower itself flowing through resistor R3 produces another source of bias voltage which also tends to oppose the original positive bias.

5. The combination of this group of voltages produces a condition whereby the grid of the cathode follower is slightly negative with respect to its own cathode and operation of the tube is on the linear portion of the characteristic curve. This operation tends to become automatic since that portion of the bias which is contributed by circuit R6-C4 is dependent upon the peak level of the synchronizing pulses.

6. The pulse output for control of scanning is taken across the un-bypassed resistor R5 in the plate of the diode sync clipper tube (T2). The circuit of Figure 97B is similar in some respects to that just described since operation of the cathode follower tube ( T1) occurs on a linear portion of the curve due to the balance of the bias developed across resistor R3 and the voltage from the plate circuit produced by the network R1 and R2. In this case, a net negative voltage of minus 2 volts appears between grid and cathode.

The cathode follower tube ( T1) performs the dual functions of video output and sync take off. The video output for the cathode of the picture tube is taken directly from the cathode load resistor (R3) through capacitor C5.

This same load resistor feeds a triode (sync separator and D-C restorer) tube T2, through the network consisting of C2, R4, R5, and C3. The combination of R5 and C3 constitutes a filter for the removal of the high frequency components of the video signal.

Sync pulses are taken from the plate of triode T2, which operates in a manner similar to that of Figure 95C. Use is made of the direct current bias voltage existing across network R6-C4 to restore the average component of the picture signal which is responsible for the average light of the televised scene.

This D-C component of the signal has been lost in passing through the video amplifier stages.

A more detailed explanation of D-C restoration will be given later in this course.

SYNC PULSE AMPLIFICATION. CLIPPING AND SHAPING. In most television receiver designs, more stages than a one tube circuit are employed for the separation of the sync pulses from the video signal. These additional stages are introduced for the purpose of: inverting the phase of the pulses (when not of the proper polarity for control of the scanning oscillator), clipping the pulse width (for more reliable control of scanning), amplifying the pulse (if it is not of sufficient strength for control), leveling the pulse (to take care of variations of the video signal and minimize the effect of interfering noise pulses). There appears to have been little standardization to date in naming these stages, and we find the following descriptive titles in the service literature of the various manufacturers: "sync clipper", "pulse stripper", "sync amplifier", "sync inverter", "sync leveler", "sync limiter", "pulse limiter" and "clamper''. These various actions are self-explanatory and have been covered in the preceding text. It should be noted that even though a stage is labeled as a "sync amplifier", it is usually biased in such a manner that either cut-off or saturation contributes some element of "leveling" or "clipping" as well as the de sired voltage amplification.

Examples of the application of these stages in typical commercial receiver designs will be found in the circuits covered under "Typical Commercial Deflection Systems" of Section 2 and in the samples of "Typical Synchronizing Systems" at the end of this section.

"SORTING" OF THE INDIVIDUAL HORIZONTAL AND VERTICAL PULSES. In the foregoing description of the various methods of separating the synchronizing pulses from the composite video signal, only the narrow horizontal pulses were mentioned. The longer vertical pulses are clipped from the signal in the same separation process.

After these sync pulses have been re moved from the video signal, by any one of the methods described, it is the n necessary to "sort" the vertical from the horizontal and feed each pulse to its respective deflection scanning systems. Since the horizontal and vertical pulses are equal in amplitude or height, the methods of separation which were described for clipping them from the video signal cannot be used to distinguish between them. They do, however, differ in time duration and it is on this basis that the ''sorting'' is accomplished.

Several times, in the description of deflection systems, mention has been made of "differentiating" networks for the removal of horizontal pulses and "integrating" networks...

Fig. 98. Horizontal pulse separation or "Differentiation".

...for the vertical pulse acceptance. At this time, we will consider the action of such systems in greater detail.

HORIZONTAL PULSE SEPARATION. The horizontal pulses of the transmitted signal are approximately five microseconds in "width" or duration as shown in Figure 65.

These pulses are impressed on a circuit of the type shown in Figure 98, which is known as a "differentiating" R-C circuit.

The term "differentiation" is a mathematical expression used in calculus and means simply the "breaking down" of a quantity into a number of small parts. In this case, the pulses of Figure 98A are "made into" smaller parts as shown in Figure 98B by the action of the circuit shown in 98C. The circuit consists of a capacitance and resistance combination in which the capacitance is in series with the separated pulse input, and the resistor is in "shunt" across the output. The time constant of this circuit is made short compared with the duration of a horizontal sync pulse. The sync pulse is held between 4 and 5 microseconds in time duration and the time constant of the horizontal differentiating circuit is made between 1 and 2 microseconds. As described here and illustrated in Figure 49, for a C-R circuit in which the time constant is short compared with the duration of the applied square wave pulse, the capacitor is completely discharged. A sharp ''pip'' of voltage occurs across the resistor at both the leading and trailing edges of the applied square wave pulse.

The amplitude of the "pip" is determined by the "steepness" of the edge of the square wave and, for this reason, limits are placed by the Federal Communication Commission on the allowable slope of the leading and trailing edges. These slopes must not occupy more than 5 percent of the horizontal line scanning interval of 63.5 microseconds.

The voltage "pip" due to the leading edge of the horizontal synchronizing square wave, is shown as a positive "pip" at (1) of Figure 98 and the "pip'' due to the trailing edge of the horizontal pulse is shown as a negative voltage at (2) of Figure 98. The leading edge pulses are the ones which are employed for control of the horizontal scanning oscillator and the negative pulses are rejected by cut-off or saturation of one or more stages of the sync system.

When the longer duration vertical synchronizing pulses arrive, the action of the differentiating circuit becomes that shown in Figure 99. Here again, a positive "pip" occurs at the leading edge of each vertical pulse and a negative "pip" at the trailing edge. The leading edge pulses continue to control the horizontal oscillator during the time of vertical retrace. In this instance, however, two pulses occur during a horizontal line-scanning interval. Only the first of these is used to control the horizontal oscillator. The second cannot cause lock-in since it happens at a time in the sawtooth wave when the amplitude is insufficient to trip the oscillator.

The horizontal pulses can be separated from the complex video signal by means other than the C -R "differentiating" circuit just described. Figure 100 shows two types of differentiating circuits which employ inductance, and a third type which uses the properties of a resonant circuit.


Fig. 99. Action of Vertical Sync Signal on Horizontal Differentiating Circuit

Fig. 100. Other Methods of Horizontal Sync Pulse Separation.




The inductance of Figure 100A is connected in series with the plate circuit of a tube which has been biased to clip the sync pulses from the video signal. The waveform of the sync-pulse plate current consists of steep slopes, which correspond to very rapid changes of current. The voltage which appears across the inductance is proportional to the rate of change of the current through it. Thus, at the leading and trailing edges of each plate current pulse, a high voltage is produced across the inductor of the same form as shown for the C -R type of differentiator (Figure 98B). If the pulses are of proper polarity and sufficient amplitude, they may be applied directly to the scanning generator by a capacitor connected to the plate end of the inductor. In case the polarity is incorrect, phase reversal may be accomplished by an amplifier stage or by transformer action as shown in Figure 100B. When a transformer is used, the secondary L2 has induced voltage pulses of opposite polarity to those occurring across the primary L1.

The secondary can be connected directly to the horizontal scanning generator grid circuit.

The circuit shown in Figure 100C operates quite differently from the two just de scribed. The resonant circuit consisting of L1 and C1 is tuned to approximately seven times the horizontal line frequency of 15,750 cycles ( 110 kilocycles). The separated sync-pulses are impressed across the circuit and "shock excite" it into oscillation at its resonant frequency. The oscillation is quickly damped out by the parallel resistor R1. Only the first half cycle of voltage across the circuit is used for control of the horizontal scanning oscillator.

This corresponds to a pulse duration of approximately 5 microseconds. Several advantages can be cited for this method of horizontal sync discrimination:

1. It allows the design of an extremely simple pulse separation and oscillator control system. The circuit can be connected directly in the plate return of the sync separator tube and can be coupled directly to the scanning oscillator because the function of pulse shaping is performed by the resonant action.

2. It is relatively immune to excitation by static or ignition noise because such pulses would have to be of the proper time duration (5 microseconds long) and of the proper repetition rate (15,750 cycles) to produce the "ringing" action. The probability of such coincidence in random noise excitation is slight. A typical example of the use of this circuit in a commercial design will be found at the end of the sync pulse separation section.

VERTICAL PULSE SEPARATION. In the description of vertical scanning systems we have mentioned "integrating" networks for segregation of the long time vertical field pulses from the" sharp" horizontal line pulses.

At this time we will consider the means of "sorting" these vertical "field scanning" pulses from the composite scanning pulses and their use to control the timing of the vertical oscillator.

Fig. 101. Vertical Pulse Separation by "Integration''

The "integrating" action which is employed to ''sort'' the vertical pulses from the complex video signal is exactly opposite in nature to the "differentiation" process used for separation of the horizontal pulses. "Integration" is a mathematical term used in calculus and means the addition of a number of small elements or parts to form a whole. Figure 101 shows the circuit employed to accomplish this action, and it will be seen that it is exactly the reverse of the "differentiation" circuit of Figure 98. In this case, the resistor is in series with the input and the capacitor is connected across the output. The time constant of the combination is much longer than that employed for "sorting" the horizontal pulses. This time constant is made approximately equal to the duration of a horizontal pulse and consequently the charge accumulated by the capacitor, due to a horizontal pulse, is s mall and will decay rapidly. This action is shown in Figure 101B. During the time shown as (1) the equalizing pulses produce only a small voltage across the capacitor. This voltage decays to zero in the interval between pulses as shown at (2). The much longer vertical synchronizing pulses produce a greater charge in the capacitor during period (3). This charge does not completely decay during the short serration interval (4). Consequently, each of the vertical pulses adds an element of charge to the capacitor and the voltage continues to build up for the entire duration of the interval of vertical pulses. The dotted line shown on Figure 101B indicates the level at which this voltage becomes of sufficient amplitude to "trigger" the vertical scanning oscillator. In most designs, this point is made to occur after two or three of the vertical pulses have charged the capacitor.

In commercial television receivers, the vertical integrating network is seldom the two element type shown in Figure 101C, but is usually a cascade network as shown in the example of Figure 101D. The resultant time constant of this network is smaller than that of any of the individual branches ( R1 -C 1, R2-C 2 or R3-C3). The value of the over-all time constant can be calculated in the same manner as used for the determination of the effect of resistors in parallel. F or the three branch circuit of Figure 101D, using T1 for the time constant R1x C1, T2 for R2xC2 and T3 for R3xC3, the effective circuit time constant (T) .l 1l 1 will be T= T1+T2+T3.

Typical values for the individual time constants for a three branch circuit such as that of Figure 101D, as found in modern receivers are from 30 to 60 microseconds. The effective over-all circuit time constants will therefore be between 10 and 20 microseconds.

The reasons for the use of cascaded integrating circuits are:

1. To prevent the erratic control of vertical retrace by random noise or static pulses. For such pulses to assume control of the vertical oscillator they would have to be comparable in duration and spacing to the vertical sync pulses.

2. To smooth out the contour of the rising voltage wave, shown in the interval 3 to 5 of Figure 101B, across the output capacitor. The action is similar to that of the familiar resistance-capacitance power supply filter system, in which the ripple is reduced by successive stages.

As a result of this smoothing action, an individual horizontal pulse cannot assume control at the instant of retrace and cause "pairing" of lines.

The sections of this cascade network are not usually made with equal time constants.

This unbalance assists in rendering the system immune from accidental triggering by noise pulses.

Fig. 102. Use of Equalizing Pulses to Assure Correctly Interlaced Vertical Scanning.

The Function of Vertical Equalizing Pulses. Here, we discussed briefly the subject of interlaced scanning which is employed to prevent "flicker" of the image. For simplicity, the retrace which occurs from bottom to top of the picture was shown at that time as a straight line or single "jump". Actually, it is necessary to keep the horizontal oscillator running in exact step or synchronism with the transmitter during the vertical retrace period which lasts from 1250 to 1400 microseconds (20 to 22 horizontal lines). Figure 102A shows a simplified version of the active downward scanning of the picture in which nine and one -half lines have been drawn to represent a field. Actually a field consists of two hundred sixty-two and one -half lines less the number of lines lost during the retrace period. The first field which starts at the upper left hand corner (Point 1) and ends at the center bottom of the picture ( Point 3) has been shown by heavy lines. The second or interlaced field, starts at the top center (Point 4) and ends at the lower right hand corner ( Point 5) and is shown by light lines. During the vertical re trace period, when the picture tube is blanked out, the beam moves upward under the combined action of both the vertical and the horizontal deflection systems . This is represented in simplified form by the diagram of Figure 102B. In this case, five and one-half lines have been used to represent the twenty to twenty-two lines actually required during the vertical retrace period. The same convention has been employed, namely, a heavy dotted line for the retrace of field No. 1 and a light dotted line for the retrace of field No. 2.

The dual functions, of producing vertical retrace at the proper instant and of keeping the horizontal oscillator in synchronism, are con trolled by the action of the groups of equalizing and vertical pulses shown in Figures 102C and 102D. It will be seen that the vertical sync signal for the retrace of field No.1 differs from that of field No. 2 by the spacing between the last horizontal pulse and the first equalizing pulse. In Figure 102C, for field No. 1, this space (a) consists of only one-half of a horizontal line since field No.1 ends at the middle of the last line as shown at Point 3 of Figure 102A. In the case of Figure 102D, for field No. 2, the space (b) between the last horizontal pulse and the first equalizing pulse consists of an entire horizontal line as shown at Point 5 of Figure 102A. Vertical blanking starts at the leading edge of the equalizing pulses and thus the successive field blanking time is accurately set up by the signal.

Even though blanking for the retrace period is accurately established, vertical retrace itself may not take place at the proper instant unless the critical charge on the integrating capacitor occurs at exactly the same point for each successive vertical sync signal. The manner in which the equalizing pulses assure this condition is shown in Figure 102E. At (1) is shown the composition of a vertical sync signal which would follow field No.1 if the equalizing pulses were not present. This signal input to the integrating circuit would charge the capacitor as shown by the dotted line labeled ( 1) in curve ( 4). This curve crosses the sync control level at time (X). The set of conditions which would occur for the type of vertical signal, without equalizing pulses, for retrace at the end of field No. 2 is shown at (2) in Figure 102E. In this case, the critical sync control level is reached at point (Y) which is sufficiently later than point (X) so that proper interlace would not occur. When equalizing pulses are employed as shown at (3), the critical "firing" point for the vertical oscillator is seen to coincide with the condition shown at (2). Successive fields preceded by equalizing pulses will therefore produce accurate control of the oscillator and assure proper interlace.

Fig. 103. Action of Horizontal Differentiation Circuits During Vertical Pulse Period

ACTION OF THE HORIZONTAL DIFFERENTIATING CIRCUIT DURING THE VERTICAL PULSE PERIOD. The formation of positive and negative "pips" respectively at the leading and trailing edges of the vertical sync pulses was described briefly on page 95.

We will now consider, in somewhat greater detail, the action of the horizontal differentiating circuit d 11ring the entire vertical pulse period. Figure 103A shows the configuration of the vertical signal which follows field No. 2.

The horizontal pulse which starts retrace of the bottom line of the picture is shown at (1) in Figure 103A and the positive output "pip" produced by its leading edge is shown at (a) in Figure 103B. The "pip" produced by the trailing edge of this same horizontal pulse, as well as those produced by the trailing edges of all of the other pulses of the period (labeled c), are rejected by the sync system as previously explained.

Each of the equalizing pulses (2, 3) which precede and follow the vertical pulse group also give rise to a pair of positive and negative pips as shown. Only the pips marked (a) are used for oscillator control. Those labeled (b) are rejected since they occur at a time in the scanning cycle at which the horizontal oscillator is not sensitive to pulse control (see Figure 68). Each of the pulses of the vertical group ( 4, 5) also produces a pair of positive and negative pips. In this case, only the positive pips (" a" of Figure 103B) are utilized.

The horizontal pulse shown as (6) is one of a group occurring during the blanking period.

The pips produced by it are identical to those produced by the horizontal pulse (1) of Figure 103.

It is evident that the vertical pulse group is able, by the action of the individual pulses and by the differences in length of these pulses, to perform the dual functions of assuring vertical retrace at the correct instant and of keeping the horizontal oscillator under control (in step with the scanning in the camera tube at the transmitter).

Vertical Pulse Separation By Use of the First Serration of the Vertical Signal. Another method of "sorting" the vertical sync pulse from the composite video signal, which depends only indirectly on R-C circuit principles, is illustrated in Figure 104. This type of vertical sync separation is not dependent, for accurate retrace and proper interlace, upon the existence of equalizing pulses.

Fig. 104. Vertical Sync Pulse Separation Using First "Serration" of Vertical Signal for Scanning Control

The first "serration" of the vertical pulse group (X1 of Figure 104A) always occurs at the same instant of time, i. e.: one-half of a horizontal line after the beginning of the first vertical pulse. This first "serration" is separated from the "mixture" and used for control of the vertical scanning oscillator. In order to accomplish this result, it must be changed from a "negative going" pulse to a "positive going" pulse. This is done by passing the separated pulse group through an amplifier stage which inverts the phase or polarity (see Figure 104B).

The inverted signal is fed to Grid No.1 of a pentagrid converter tube (6SA7), see Figure 104E, through a differentiating network (C1- R1). This circuit has a time constant of approximately one-half of the horizontal line scanning time or about 30 microseconds. This grid is biased negatively to such a value that the differentiated pulses, due to the equalizing pulse period, are not of sufficient amplitude to cause any flow of plate current (see Figure 104C).

When the inverted pulse arrives, due to the first serration of the vertical signal, (Point X3 of Figure 104C) it has developed sufficient voltage, across R1, to cause the grid to pass its cut-off bias and start a pulse of plate current.

Screen Grids Nos. 2 and 4 are connected together and are held at a low voltage by the high resistance divider network (R2, R3). Bypass capacitor C2 is of relatively small value and as Grid No.1 is driven positive by pulse X3 the screen voltage drops rapidly.

The third grid, normally the input when the tube is acting as a converter, is coupled to the screens through capacitor C3. This grid will undergo the same voltage drop as the screens and will cut off the plate current as the screen voltage goes negative. Grid No. 3 is held beyond cut-off until the charge on capacitor C3 has "leaked" off through resistor R4. The time constant of C3-R4 is made sufficiently long to allow the rest of the vertical pulse group to pass before Grid No. 3 can regain control.

As the result of this action, only the first pulse which passes the cut-off bias of Grid No. 1 (Point X3) will cause any plate current. This plate current occurs as a single sharp "pip" pulse is coupled to the vertical scanning oscillator through capacitor C4.

SYNCHRONIZING SYSTEMS (METHODS OF UTILIZING THE SEPARATED SYNC PULSES): There are three general types of synchronizing systems in use in modern television receivers.

They are classified as follows:

1. "Triggered" sync. This system depends upon the use of individual sync pulses to control the scanning of each horizontal line and the vertical retrace of each field. The elements of this type of system have been covered in the preceding text and the circuit illustrations of pages 74 and 84.

2. "Flywheel" or A. F. C. sync. The A. F. C. system, which was described briefly on pages 75-76 and illustrated in Figure 83, depends upon the average of the synchronizing pulses and their rate of repetition rather than the individual pulses. A fully detailed description of the circuit operation will be presented in this section.

3. "Pulse width" controlled sync. In this method, the sync pulse is compared in timing with the output of the scanning oscillator and produces a narrow, augmented pulse for control of the instant of retrace. Its action has been covered on pages 83-84 and illustrated by Figure 90.

Fig. 105. "Flywheel" or A. F. C. Control of Horizontal Scanning Oscillator


Figure 105 shows a typical circuit adaptation of the automatic frequency control of horizontal scanning by the sync pulses. More properly, it should be called "automatic phase control" since its action "locks" the phase, as well as the frequency, of the horizontal oscillator.

The operation of the circuit is as follows:

1. A very stable oscillator, tube T4 and its associated circuits, generates a sine wave of 15,750 cycles per second. This oscillator circuit is of the "hot cathode" Hartley type and its frequency determining components are inductance L3 and capacitance C2. The free running frequency is adjusted by the powdered iron core of inductor L3 with the hold control (R7) in mid-position.

2. A tapped coil circuit , L 1-L2 , is tightly coupled to L3 and tuned slightly "off resonance". The voltage across these coils is applied to a ''discriminator" circuit similar to the type used for frequency modulation detect ion. The voltages, from the horizontal oscillator, which are applied to the plates of the discriminator tubes ( T1 and T2) are equal in amplitude and opposite in phase (180 degr. out of phase with each other). See Figure 106, (1) and (2).

3. The "clipped" and amplified sync pulses are applied to the discriminator across resistor R3 and appear "in phase" and of the same amplitude at each diode of the discriminator as shown in Figure 106.

4. Three conditions of the system in which the relation of the oscillator frequency to the sync pulse repetition rate is slow, correct, and fast are shown in Figures 106A, 106B, and 106C respectively. It will be noted that the diode load resistors R1 and R2 are so connected that the voltage from ground to point X is the difference between the rectified output of the diode, tubes T1 and T2. Arrows show the direction of electron flow due to the plate current of these tubes.

5. if tube T1 alone is conducting, point X will be positive with respect to ground and if tube T2 is conducting, with no current flow through tube T1, point X will be negative with respect to ground. In the abs enc e of sync pulses, equal and opposite voltages appear across the diodes, tubes T1 and T2, and the total rectified voltage appearing between point X and ground is zero, over the cycle.

6. The condition of stable operation occurs with the sync pulses riding at zero phase, or on the axis of the wave as shown at (B) of Figure 106. In this case, the rectified output of the upper diode, tube T1, would pro duce a voltage across R1, as shown in Figure 106B (3). The lower diode, tube T2, would pro duce the voltage wave shown in Figure 106B (4). These voltages oppose each other due to the method of connection of the diodes to the series load resistors, R1 and R2. The net charge on the filter network, C3, R4, and C4; is zero, as shown in Figure 106B (5).

7. If the horizontal scanning oscillator is running faster than the repetition rate of the sync pulses (see Figure 106A), the upper diode output voltage will exceed that of the lower diode as shown at 106A (3). After smoothing through the filter network, a positive voltage will appear at the grid of reactance control tube (T3).

8. In a similar manner, if the oscillator is running slower than the repetition rate of the sync pulses, the set of conditions shown in Figure 106C will apply. In this case, a negative control voltage will appear at the grid of tube T3.

Fig. 106. Discriminator Action in Horizontal AFC Sync Control Circuit of Figure 105

Fig. 107. Reactance Control Operation

9. The 6AC 7 reactance control tube is connected across the oscillator "tank" inductance L3. The manner in which this tube can control the instantaneous frequency of the oscillator is as follows:

a. The plate current versus plate voltage curves of a high gain, sharp-cut-off pentode, such as the 6AC7, exhibit a long range of plate voltage over which there is substantially no change of plate current. This region is shown in Figure 107B. The plate current is under the control of the grid voltage as indicated by the individual curves of the plate current-plate voltage family.

b. Referring to Figure 107 A, we find that the plate is connected to the high side of the oscillator tuned circuit through capacitor C2. The alternating voltage from the oscillator. appearing across the tube, is swinging over the flat portion of the curve and therefore no change of plate current occurs. If, on the other hand, an alternating volt age is applied between grid and cathode the plate current will be changed in amplitude, and this change will be in phase with the alternating grid voltage.

c. If the alternating voltages applied to the grid and plate are made 90° out of phase, the plate current will then have an alternating component which is 90° out of phase with the plate voltage, i. e. in phase with the grid voltage.

d. Any circuit in which the current flowing is not in phase with the applied voltage is reactive. The circuit then appears to be capacitive or inductive.

Thus, tube T1 of Figure 107A can be made to appear as a reactance connected across the oscillator inductance L 1.

e. The phase shift of applied alternating grid voltage, with respect to the alternating plate voltage, is provided by the network C1- R1. The voltage across R1 is leading the voltage across the tank circuit by almost 90°. Since the reactance of the coupling capacitor C2 is low, the voltage applied between cathode and grid is approximately 90 degr., out of phase with that applied between plate and ground.

f. The control grid is returned to ground, or to the end of R1 opposite the cathode, through capacitor C 3. The reactance of C3 is very low at the oscillator frequency of 15,750 cycles.

The DC return path of the control grid is through the output load resistors of the discriminator stage ( R1 and R2 of Figure 105). Since the alternating grid voltage is applied between cathode and the grounded grid, rather than between grid and ground, the plate differs by 150° from the phase it would have if the cathode had been grounded and the grid voltage made variable. In this case, the plate current will lag the plate voltage and the tube will appear as an inductance to the tuned circuit.

g. The bias voltage on the grid of the reactance tube controls the value of its transconductance (ratio of change in plate current to the change in grid voltage which caused it). The alternating plate current amplitude can thus be changed by varying the bias on the control grid. A low or positive bias will result in an increase in the AC plate current while a more negative bias will reduce the value of AC plate current. Since the AC plate voltage is fixed by the output of the oscillator, a bias change will change the ratio of voltage to current and thus, the apparent value of inductance across the oscillator circuit.

10. If the frequency of the horizontal oscillator should shift with respect to the sync pulse rate, due to variation of supply voltage or other cause, a change of the DC output of the discriminator would occur. This changes the transconductance of the control tube, shifting the instantaneous oscillator frequency to restore the equilibrium condition of Figure 106B. The network of resistors and capacitors, (R4, C3 or C3A, C4 and R5), which connect the output of the discriminator to the grid of the control tube T3 (Figure 105) requires explanation. The purpose of resistor R4 is to provide a DC return path for the grid of the control tube (T3). Resistor RS acts as a parasitic suppressor to prevent any possible ultra-high frequency oscillation which might occur in the high mutual conductance type 6AC7 control tube.

Capacitor C3 and capacitor C4 constitute a voltage divider and a filter. The "ripple" on the rectified output of the discriminator diodes is smoothed to furnish a DC voltage to the control tube. Since the reactance of the smaller capacitor C3 (.004 mfd) is on the top side of the divider, and the larger capacitor C4 (.05 mfd) on the bottom, extremely rapid volt age changes are prevented from affecting the grid voltage of the control tube. Rapid changes can be caused by bursts of noise or by the serrations of the vertical sync signal. These changes are rejected by the action of this volt age divider.

The ratio of the voltage divider consisting of C3 and C4 can be changed by "switching'' capacitor C3A (.01 mfd) in parallel with C3. This link connection for the addition of C3A is provided to take care of a condition existing in some television transmitters. If any "phase" modulation of the exact position of the horizontal sync pulse occurs, it is necessary for the A. F. C. control tube to follow this modulation and a "faster" response circuit is required. This is provided by increasing the capacitance of C3 by the addition of C3A. The effect of the phase modulation of the horizontal pulses, if suppressed by the original capacitance ratio, would be to produce a horizontal displacement of part of the picture with respect to the raster.

The use of the higher capacitance, C3A, will result in some sacrifice of "immunity" of the system from horizontal "tearing" by noise. Transmitting technique is constantly improving and the service technician should re-check any installation in which it was originally necessary to employ the higher capacitance to determine whether the advantages of a "fast" circuit with its attendant noise immunity can be tolerated.

The A. F. C. type of sync system has been covered in detail because its use is becoming increasingly popular in television receiver de sign. The reasons for its popularity are: (1) its relative immunity from triggering by noise, (2) it can take advantage of the entire horizontal pulse time for the discriminator control, and ( 3) it allows the horizontal trace return time to be slower than could be tolerated with "triggered" sync operation. The reason for its immunity to the effects of static, man-made interference, and receiver input-circuit noise lies in the fact that the frequency distribution of such noise energy is approximately the same both above and below the sine wave oscillator frequency. Thus, the average DC contribution of the two discriminator diodes due to the noise tends to be equal and cancel at point "x" of Figure 105.

Fig. 108. Composite Circuit Showing Points at which Sync Pulses Might Be Separated

TYPICAL COMMERCIAL SYNC SYSTEMS. The circuits described and illustrated on pages 73 to 84 covered the sync-controlled scanning system from the point of separation of the pulses to the deflection of the electron beam in the picture tube. Means of separating the sync pulses from the complex video signal have been covered on pages 85 through 98. It has been shown that sync pulse separation can be accomplished at several places in the circuit between the video detector tube and the picture tube.

To illustrate the many points in the television receiver system at which the sync synchronizing pulses may be separated from the signal, a composite circuit is presented in Figure 108. It should be understood that this circuit does not represent an actual or practical television system since extra tubes have been added to present the possible points of sync separation. Four possible positions of sync separation are shown at "A", "B", "C", and "D". The choice of any one of these points by the receiver designer is governed by the required polarity of the pulses and by the amount of signal energy required by the remainder of the sync system.

When a diode such as tube T3 is coupled directly to the output of the video I-F amplifier, illustrated by sync separator No.1 of Figure 108, the pulses obtained across the cathode resistor R3 are positive with respect to ground.

If sync separation is taken at this point, it is usually necessary to employ additional amplification to obtain a pulse of sufficient amplitude to control the horizontal and vertical scanning oscillators. Sync separation at the video I-F output has the advantage of being independent from the setting of the video gain or "contrast control" R1. When separation is done at this point the last I-F transformer secondary, L-2, is usually phased so that the negative half of the video carrier on the last IF amplifier grid is rectified and employed for sync control.

This is done to take advantage of the fact that strong noise impulses will be "clipped", to some extent, since they are more negative than the sync pulses and will drive the last IF amplifier tube beyond plate cut-off, thus causing it to become a "limiter". The action of tube T3 at this point has been covered earlier and illustrated in Figure 93.

Sync signal separation at point "B" occurs after rectification of the video carrier and since the rectified video has passed through the amplifier tube T4, the polarity of the signal will be negative as shown by the wave form at point "B" of Figure 108. A signal taken at this point is usually passed through an inverter stage to reverse its polarity and then applied to a sync separator (the triode type covered by Figure 95A or the pentode type of Figure 96). Separation of the pulse by tubes T5 or T6, as shown at "C" and "D", occur at high level and require very little, if any, additional amplification. When the method represented by tube T5 is used, the tube fulfills only the function of sync separation. The pulse at "C" is negative with respect to ground and can be used for the direct control of cathode-coupled multivibrators after differentiation or integration.

Fig. 109. Horizontal Sync Pulse Differentiation by Means of a Tuned Circuit. (Also See Figure 100C)

When diode tube T6 is employed , it serves the dual function of sync separation and DC restoration. Restoration of the direct current component of the signal is required since the use of capacitance-coupled video amplifiers causes the loss of DC component present in the video signal. The necessity for restoration of the DC component, which is responsible for the average background lighting of the scene, will be covered when we consider the video signal in greater detail.

It should be noted that any particular television receiver circuit will employ only one of the sync separation tubes shown, and that the circuit of Figure 108 would reduce to four tubes rather than the six tubes shown. One exception to this reduction exists when the horizontal pulses are taken off by one separator, such as tube T3, and the vertical pulses at a later tube, such as T5 or T6.

Figure 109 shows an unusual sync separation system in which a single type 12SN7 tube is employed for the functions of video output, DC restoration and sync separation. In this case, the second section of tube T1 acts as a "kangaroo" cathode follower. The plate load of this tube, resistor R1, develops the sync pulses. The long time constant circuit comprised of R2 and C2 develop the direct current bias required for restoration of the background lighting level. Separation of the vertical sync pulses is accomplished by coupling the vertical oscillator tube T2 directly to the plate of tube T1 through the integrating network consisting of C3, R5, C4, R6, and C5.

Horizontal pulse differentiation is accomplished by the use of a tuned circuit comprised of inductor L 1and its associated circuit capacitances. This circuit is resonant at a frequency of approximately 110 kilocycles and is allowed to oscillate for one half-cycle or 5 microseconds. This us e of resonance was covered in the text of page 96 and illustrated in Figure 100. Since this first half-cycle pulse controls a cathode-coupled multi vibrator, it must be a "negative-going" pulse as discussed under the description of pulse control. The input section of tube T3 has a net bias determined by its cathode resistor R7 a,1d the divider from the plate circuit of tube 'fl.

This plate circuit network consists of R1, R3 and the parallel resistance of L1 and R4.

Resistor R4 "damps" the shock-excited oscillation of the tuned circuit, caused by the horizontal sync pulses. This damping prevents the continuance of oscillation beyond the first half-cycle, which is required for the control of the horizontal scanning oscillator tube T3.

SUMMARY OF SYNCHRONIZATION METHODS AND SYSTEMS: As explained in the introduction to this course, the major difference be tween television receivers and other types of radio equipment, is the group of circuits which are peculiar to the formation of a cathode-ray produced image, and its control by scanning systems at the transmitter and receiver. The horizontal and vertical scanning systems, the use of pulses as part of the television carrier for control of scanning, and the methods of segregating or separating these pulses have been covered in considerable detail up to this point. Experience with television receivers, which have been in service for the past several years in metropolitan areas, discloses the interesting fact that the majority of service failures have occurred in the scanning and synchronization circuits of these sets. In many cases, the trouble has been readily traced to vacuum tube failures, especially in multivibrator and blocking oscillator circuits. Ageing of components of these oscillator circuits has quite often been responsible for a drift of frequency. This is usually serious enough to cause failure of the receiver scanning· to "lock-in" with the transmitted signal. The service technician is urged to study these portions of the television system thoroughly, since the proper installation of new receivers and the maintenance of those in service, will, in the majority of cases, require adjustment or re pair of scanning and synchronization circuits.

Fig. 110. Control Characteristic of 10BP4 Picture Tube

BRIGHTNESS CONTROL: Our previous study of the television receiver system has concerned itself with the motion of the electron beam in the picture tube to accomplish scanning of the picture area. It was shown in Section 1, that the intensity of illumination of the cathode -ray screen surface could be controlled by a grid cylinder in the picture tube "gun" structure. We have indicated at several points in the text that the variation of video or picture signal on this control grid is responsible for the instantaneous changes of spot illumination which make up the elements of the picture. The signal voltage on this control element is negative in nature so that a swing in the negative direction produces a darker spot. Finally, at some critical negative voltage, the spot of light is entirely extinguished.

Figure 110 shows the relation of the control grid voltage to the brightness of the picture spot for a typical ten inch picture tube (Type 10BP4). In this particular tube, with normal voltages on other elements, the spot is extinguished (picture tube black} for all grid voltages below a value of positive five volts. A swing to a positive 35 volts on the control grid, with respect to the cathode, produces the brightest spot or "highlight" of the picture.

One of the essential controls of the television receiver is an adjustment of the bias on the grid to assure that the blanking level or pedestal of the signal occurs at the "black" point which, in the case of this particular tube, would be 5 volts. Figure 110B shows a bias system in which the voltage, E, established by the adjustment of potentiometer R1 biases control grid G1 with respect to cathode and determines the correct picture brightness. In this case, the polarity of the video signal is such that the plate of the video output tube is connected to the control grid. It is possible to operate the picture tube with the control grid connected to the low side of the video output stage and the cathode coupled to the high or plate side, as shown in Figure ll0C. In either case, the "brightness" control is a voltage adjustment of the bias between the control grid and cathode, which establishes the correct blanking or black level.

As the "brightness" control is adjusted, with no picture signal present, a point will be reached at which the vertical retrace lines, caused by the motion of the spot from bottom to top of the picture, will become visible. The control should be adjusted in a counter clock wise direction until these lines just disappear.

The "brightness" control, as well as the "contrast" control, are normally made "front panel" or customer operated, to allow adjustment of picture quality to accommodate the existing conditions of room illumination.

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Updated: Tuesday, 2021-11-16 17:56 PST